This invention relates to an apparatus for detecting fuel dielectric constant which detects the dielectric constant of a fuel in a non-contact mode which is supplied to a burner or the like to determine the properties of the fuel, and more particularly relating to an apparatus to detect fuel dielectric constant for measuring the content of alcohol in a fuel used for the engine of a motor vehicle or the like.
Recently, in the United States of America and in many countries in Europe, in order to decrease the consumption of oil and to reduce the air pollution due to the exhaust gas of motor vehicles, a fuel prepared by mixing alcohol with gasoline is being introduced for motor vehicles. However, in case of using the alcohol-mixed fuel to the engine which is adjusted an air/fuel ratio to match with a gasoline fuel, since alcohol is smaller in theoretical air/fuel ratio than gasoline, the air/fuel ratio is leaned, which makes it difficult to smoothly operate the engine. In order to eliminate this difficulty, the following method has been employed: The content of alcohol in the alcohol-mixed fuel is detected, and the air/fuel ratio and the ignition timing are adjusted according to the content of alcohol thus detected.
In order to detect the content of alcohol, there have been proposed a method in which the dielectric constant of an alcohol-mixed fuel is detected, and a method in which the refractive index of the same is detected. With respect to the former method, the present Applicant has proposed a "dielectric constant detecting apparatus" under Japanese Patent Application No. 22488/1991.
The dielectric constant detecting apparatus thus proposed will be described with reference to FIG. 5.
In FIG. 5, a sensor section A is provided. And a bottomed cylindrical insulating tube 1 is made of an insulating material such as ceramic or oil-resisting plastics into which a fuel is led. An electrically conductive electrode 3 in the form of a cylinder is provided inside the insulating tube 1 in such a manner that it is coaxial with the insulating tube 1 with its outer cylindrical surface being substantially in parallel with the inner cylindrical surface of the insulating tube 1. A single layer coil 4 is wound on the insulating tube 1 in such a manner as to confront with the electrically conductive electrode 3. The lead wires 4a and 4b are provided to the single layer coil 4. A fuel passageway 2 is defined by the inner cylindrical surface of the single layer coil 4 which contacts with the insulating tube 1 and the outer cylindrical surface of the electrode 3. A flange 5 is provided with the electrode 3, and is coupled to the insulating tube 1 through a fuel seal 7 so as to form a fuel container (in this case, the flange 3 being integral with the electrode 3). Nipples 6 lead the fuel to the fuel passageway 2. A detecting circuit section B.
The detecting circuit section B comprises: a series resistor 10 (whose resistance being R.sub.s) connected in series to the lead wire 4a of the single layer coil 4; a 0.degree. phase comparator connected in parallel to the resistor 10; a low-pass filter connected to the output of the phase comparator 14; a comparison integrator 16 which is connected to the output of the low-pass filter 15 and to which a predetermined reference voltage V.sub.ref corresponding to a phase shift 0.degree. is applied; a voltage-controlled oscillator 17 connected to the output of the comparison integrator 16; an amplifier 18 for amplifying the output of the voltage-controlled oscillator 17; and a frequency divider 19 adapted to frequency-divide the output of the voltage-controlled oscillator 17.
The operation of the above conventional apparatus to detect dielectric constant will be described.
The sensor section A is arranged as shown in FIGS. 4(a) and 4(b). In FIGS. 4(a) and 4(b), the inductance L of the single layer coil 4 is contained. The capacitance C.sub.f between the single layer coil 4 and the electrically conductive electrode 3 changes according to a change of the dielectric constant .epsilon. of the fuel in the fuel passageway 2. And a capacitance C.sub.p is as a stray capacitance of the lead wire 4a, an input capacitance of the phase comparator 11 and the like, which is not effected by the dielectric constant .epsilon..
Hereupon, when the frequency applied to the lead 4a of the sensor section A is changed, a parallel LC resonance occurs as shown in FIG. 4(c). In this case, the parallel resonance frequency f.sub.r can be calculated according to the following Equation (1): ##EQU1##
where K, a and b are the constants which are determined according to the configuration of the sensor section A. As is seen from Equation (1), the resonance frequency f.sub.r depends on the dielectric constant .epsilon. of the fuel; that is, as the dielectric constant .epsilon. increases, the resonance frequency f.sub.r is decreased.
The resonance frequency f.sub.r of a concrete example of the sensor section having a predetermined configuration was measured as follows: In the case where the fuel was methanol having a dielectric constant .epsilon.=33, the resonance frequency f.sub.r was 7.5 MHz; and in the case where it was gasoline having a dielectric constant .epsilon.=2, the resonance frequency was about 9.5 MHz. In the case where a fuel was prepared by mixing methanol and gasoline in an optional mixing ratio, the resonance frequency f.sub.r changed according to the content of methanol as shown in FIG. 4(d). Hence, by detecting a signal corresponding to the resonance frequency f.sub.r, the dielectric constant .epsilon. of the fuel, and accordingly the content of methanol in the methanol-mixed fuel can be detected.
The detecting circuit section B, designed to detect the resonance frequency f.sub.r, operates as follows:
With a methanol-mixed fuel in the fuel passageway 2, the amplifier 18 applies a high frequency signal to a series circuit of the resistor 10 and the single layer coil 4. The voltage signal across the resistor 10; i.e., a high frequency voltage signal applied to the series circuit, and a high frequency voltage signal applied to the single layer coil 4 are applied to the phase comparator 14, where their phases are compared with each other.
It is assumed that the frequency of the high frequency voltage signal applied to the series circuit is equal to the resonance frequency f.sub.r. In this case, as shown in FIG. 4(c), the current voltage phase of the sensor section A is 0.degree., and therefore the phase shift between the high frequency voltage signals provided at both ends of the resistor 10 is also 0.degree.. When, on the other hand, a high frequency voltage signal whose frequency is lower than the resonance frequency f.sub.r as shown in FIG. 4(c) the current voltage phase of the sensor section A leads 0.degree., and therefore with the phase of the high frequency signal applied to the series circuit as a reference, the phase shift between the high frequency voltage signals provided at both ends of the resistor 10 is larger than 0.degree..
Thus, a phase synchronization loop is established in which the output of the phase comparator 14 is converted into a DC voltage corresponding to the phase shift with the aid of the low-pass filter 15; this DC voltage and the DC voltage V.sub.ref corresponding to a phase shift 0.degree. are applied to the comparison integrator 16, where a difference between the phase shifts is subjected to integration; and the output of the comparison integrator 16 is applied to the voltage-controlled oscillator 17 which applies the high frequency signal through the resistor 10 to the above-described series circuit.
With the phase synchronization loop thus established, the voltage-controlled oscillator 17 is so operated that the phase shift between the high frequency voltage signals at both ends of the resistor 10 be 0.degree., and the oscillator 17 oscillates at the resonance frequency f.sub.r at all times. The frequency divider 19 subjects the output frequency of the voltage-controlled oscillator to frequency division to provide a frequency output f.sub.out. Since the oscillation frequency of the voltage-controlled oscillator 17 corresponds to the control input voltage in a ratio of 1:1, the output of the comparison integrator 16 can be used as a voltage output V.sub.out.
The conventional dielectric constant detecting apparatus will be described more concretely with reference to FIGS. 6 and 7. As shown in FIG. 6, the phase comparator 14 includes an EXCLUSIVE OR circuit 14c, and the phase synchronization loop is so formed that the phase shift between the high frequency voltage signals at both ends of the resistor 10 be 0.degree.. FIG. 7 shows signals P1 through P6 at various circuit points in FIG. 6. The signal P1, or a high frequency square wave signal P1 outputted by the voltage-controlled oscillator 17 is applied to the CK port of a first D flip-flop circuit 18 in the amplifier 18, and it is further applied through an inverter 18c to the CK portion of a second D flip-flop circuit 18b with its phase inverted A signal at the inversion output port of the first D flip-flop circuit 18a is applied to the D port of the second D flip-flop circuit 18b, and a signal at the inversion output port of the second D flip-flop circuit 18b is applied to the D portion of the first flip-flop circuit 18a. The signal P2 is provided at the output port Q of the first D flip-flop circuit 18a, being the high frequency signal applied to the single layer coil 4 through the resistor 10. The signal P2 is changed in level at the rise of the high frequency square wave signal P1; that is, the signal P2 corresponds to a signal obtained by subjecting the signal P1 to 1/2 signal frequency division. The signal P2 is applied through an inverter 14a to the EXCLUSIVE OR circuit 14c. On the other hand, the signal P3 is provided at the output port Q of the second D flip-flop circuit 18b, and it is changed in level at the fall of the signal P1; that is, the signal P3 is equal in frequency to the signal P2 and different by 90.degree. in phase from the latter P2.
The signal P4 is provided in the connecting line between the resistor 10 and the single layer coil 4 so that it is applied to the latter 4. The signal P4 is further applied to one input terminal of the EXCLUSIVE OR circuit 14c, while the signal P3 is applied through an inverter 14b to the other input terminal with its phase inverted, so that those signals are subjected to phase comparison. The high frequency signal P4 provided in the connecting line between the resistor 10 and the single layer coil 4 is sinusoidal as shown in FIG. 7. There-fore, the DC level of the signal P4 is adjusted to the threshold level of the inverter 14a by an operational amplifier 20 with the aid of a variable resistor 21; that is, the sinusoidal signal P4 is shaped into the signal P5 which is a square wave.
At the frequency at which the LC circuit of the sensor section A resonates, the output square wave P4 of the inverter 14a is opposite in phase to the square wave P2 applied to the resistor 10, and its phase is shifted by 90.degree. from that of the signal P3 at the output port Q of the second flip-flop circuit 18b. Therefore, when the phase shift between the signals P2 and P4 at both ends of the resistor 10 is 0.degree.; that is, when the frequency provided is the one at which the LC circuit of the sensor section A resonates, the output of the EXCLUSIVE OR circuit 14c, namely, the signal P6 is a square wave having a duty of 50%. When the frequency is other than the resonance frequency, the duty of the signal P is smaller than or larger than 50%. That is, the square wave provided by the EXCLUSIVE OR circuit has the duty corresponding to the phase shift between the signals P2 and P4 in a ratio of 1:1.
The output signal P6 of the EXCLUSIVE OR circuit 14c is applied to the low-pass filter 15, the DC output of which corresponds to the phase shift between the high frequency voltage signals P2 and P4 at both ends of the resistor 10 in a ratio of 1:1. The output of the low-pass filter 15 is applied to the comparison integrator 16, where the shift between it (the output) and the voltage V.sub.ref is subjected to integration. It should be noted that the voltage V.sub.ref has been so adjusted with a variable resistor 22 that it is equal to the DC level which the low-pass filter 15 outputs when the phase shift between the signals P2 and P4 at both ends of the resistor 10 is 0.degree.. The resultant integration value; i.e., the output of the comparison integrator is applied to the voltage-controlled oscillator 17 to control the oscillation frequency.
That is, the phase synchronization loop formed controls the output frequency of the voltage-controlled oscillator 17 so that the phase shift between the high frequency voltage signals at both ends of the resistor 10 be 0. Hence, the frequency output f.sub.out which is obtained by frequency-dividing the output frequency of the voltage-controlled oscillator 17 is a function which decreases monotonically with respect to the fuel dielectric constant .epsilon. shown in FIG. 4; i.e., the methanol content. The output of the comparison integrator applied to the voltage-controlled oscillator 17 is outputted as a voltage output V.sub.out.
The conventional dielectric constant detecting apparatus thus organized is disadvantageous in the following points:
When the fuel dielectric constant changes abruptly so that the phase synchronization loop becomes unsatisfactory in control, the signals P2 and P4 become different in phase, and the impedance of the LC resonance circuit is decreased as shown in FIG. 4, and the threshold level of the inverter 14a adapted to shape the waveform of the sinusoidal high frequency signal P4 differs somewhat from the DC level which is applied to the single layer coil 4 with the aid of the operational amplifier 20 and the variable resistor 21. Hence, as shown in FIG. 8, the signal P4 no longer crosses the threshold level, as a result of which no waveform shaping operation is carried out as indicated at P5 in FIG. 8.
In this case, the output P6 of the phase comparator 14 is a signal having a duty of 50% which is obtained merely by inverting the signal P3, and the output of the low-pass filter 15 is the same as that which is provided when control is made by the phase synchronization loop. As a result, the phase synchronization fails in control, so that a value different from the true fuel dielectric constant is outputted.
In the case where the conventional fuel dielectric constant detecting apparatus is manufactured on mass-production, the DC levels to be applied to the single layer coils 4 must be adjusted individually according to the threshold levels of the inverters 14a.
Furthermore, when the duty of the output of the voltage-controlled oscillator 17 is not 50%, or the supply voltage applied to the detecting circuit B changes, the high level voltage of the EXCLUSIVE OR circuit 14c is changed, and the output of the low-pass filter 15 is changed which is the DC signal corresponding to the phase shift between the voltage signals at both ends of the resistor 10, so that, when the phase shift is 0.degree., the DC level voltage is changed. This means that the aimed phase shift of the phase synchronization loop is shifted from 0.degree.. That is, as shown in FIG. 9, a frequency f.sub.0 should be outputted; however, a frequency f.sub.1 is outputted because the aimed phase shift is shifted as was described above. On the other hand, as the fuel dielectric constant changes, the resonance is changed in quality factor Q. For instance, as indicated by the two-dot chain lines in FIG. 9, the phase curve becomes gentle in inclination with the quality factor Q decreased, and a frequency f.sub.2 is outputted. In this case, the detection is low in accuracy, and affected by the fuel conductivity.
In a mass production of the conventional dielectric constant detecting apparatus, for each comparison integrator 16 the voltage V.sub.ref must be adjusted according to the duty of the output of the voltage-controlled oscillator 17. This will lower the manufacturing efficiency.